Switchable Tunable Acoustic Resonator Using BST Material

ABSTRACT

An acoustic resonator includes a first electrode, a second electrode, and a barium strontium titanate (BST) dielectric layer disposed between the first electrode and the second electrode, where the acoustic resonator is switched on as a resonator with a resonant frequency if a DC (direct current) bias voltage is applied across the BST dielectric layer. The acoustic resonator is also switched off if no DC bias voltage is applied across the BST dielectric layer. Furthermore, the resonant frequency of the acoustic resonator can be tuned based on a level of the DC bias voltage, with the resonant frequency increasing as the level of the DC bias voltage applied to the BST acoustic resonator increases.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation application of, and claims priorityunder 35 U.S.C. §120 from, co-pending U.S. patent application Ser. No.11/682,252 entitled “Switchable Tunable Acoustic Resonator Using BSTMaterial,” filed on Mar. 5, 2007, which claims priority under 35 U.S.C.§119(e) from U.S. Provisional Patent Application No. 60/780,229,entitled “Piezoelectric Switch and FBAR,” filed on Mar. 7, 2006, andfrom U.S. Provisional Patent Application No. 60/835,253, entitled“Voltage-Controlled Film Bulk Acoustic Resonators,” filed on Aug. 2,2006, the subject matters of all of which are incorporated by referenceherein in their entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention generally relates to acoustic resonators and theirapplications in electronic circuits.

2. Description of the Related Art

Capacitors are a basic building block for electronic circuits. Onedesign for capacitors is the parallel-plate configuration, in which adielectric is sandwiched between two electrodes. FIG. 1 is a blockdiagram illustrating a typical metal-insulator-metal (MIM) parallelplate configuration of a thin film capacitor 100. The capacitor 100 isformed as a vertical stack comprised of a metal base electrode 110 bsupported by a substrate 130, a dielectric 120, and metal top electrode110 a. The lateral dimensions, along with the dielectric constant andthickness of the dielectric 120, determine the capacitance value.

Materials in the barium strontium titanate (BST) family havecharacteristics that are well suited for use as the dielectric 120 insuch capacitors 100. BST generally has a high dielectric constant sothat large capacitances can be realized in a relatively small area.Furthermore, BST has a permittivity that depends on the applied electricfield. In other words, thin-film BST has the remarkable property thatthe dielectric constant can be changed appreciably by an appliedDC-field, allowing for very simple voltage-variable capacitors(varactors), with the added flexibility that their capacitance can betuned by changing a bias voltage across the capacitor. In addition, thebias voltage typically can be applied in either direction across a BSTcapacitor since the film permittivity is generally symmetric about zerobias. That is, BST typically does not exhibit a preferred direction forthe electric field. One further advantage is that the electricalcurrents that flow through BST capacitors are relatively small comparedto other types of semiconductor varactors.

Such electrical characteristics of the BST capacitors allow otherpotential beneficial uses of the BST capacitors in electronic circuits.

SUMMARY OF THE INVENTION

Embodiments of the present invention include an acoustic resonatorcomprising a first electrode, a second electrode, and a barium strontiumtitanate (BST) dielectric layer disposed between the first electrode andthe second electrode, where the acoustic resonator is switched on as aresonator with a resonant frequency if a DC (direct current) biasvoltage is applied across the BST dielectric layer. The acousticresonator is also switched off if no DC bias voltage is applied acrossthe BST dielectric layer. Furthermore, the resonant frequency of theacoustic resonator can be tuned based on a level of the DC bias voltage,with the resonant frequency increasing as the level of the DC biasvoltage increases.

In one embodiment, the acoustic resonator is formed on a sapphiresubstrate. In another embodiment, the acoustic resonator is formed overan air gap disposed between the second electrode and a substrate. Instill another embodiment, the acoustic resonator is formed over anacoustic reflector disposed between the second electrode and asubstrate, where the acoustic reflector is comprised of a plurality ofalternating layers of platinum (Pt) and silicon dioxide (SiO₂) andreduces damping of the resonance of the acoustic resonator caused by thesubstrate. In still another embodiment of the present invention, theacoustic resonator includes a first part formed on a substrate and asecond part formed over an air gap.

The BST based acoustic resonator of the present invention functions canbe switched on or off simply based on whether a DC bias voltage isapplied or not, and its resonant frequency can be tuned based on thelevel of the DC bias voltage. Thus, the BST based acoustic resonator hasmany versatile uses in electronic circuits, such as switchable, tunablefilters and a duplexer for transmitting and receiving a radio frequencysignal over an antenna.

BRIEF DESCRIPTION OF THE DRAWINGS

The teachings of the embodiments of the present invention can be readilyunderstood by considering the following detailed description inconjunction with the accompanying drawings.

FIG. 1 is a block diagram illustrating a typical metal-insulator-metal(MIM) parallel plate configuration of a thin film BST capacitoraccording to one embodiment of the present invention.

FIG. 2 is a graph illustrating RF transmission measurements of the BSTcapacitor of FIG. 1 as a function of the frequency of the RF signal.

FIG. 3 is a graph illustrating RF transmission measurements of the BSTcapacitor of FIG. 1 as a function of the frequency of the RF signalunder different DC bias voltages.

FIG. 4 is a diagram of an equivalent circuit modeling a piezo-electrictransducer.

FIG. 5 illustrates the structure and use of the BST-based FBAR (FilmBulk Acoustic Resonator) according to one embodiment of the presentinvention.

FIG. 6 illustrates the structure of the BST-based FBAR, according to oneembodiment of the present invention.

FIG. 7A illustrates the structure of the BST-based FBAR, according toanother embodiment of the present invention.

FIG. 7B illustrates the various structures of the acoustic reflectorthat can be used with the BST-based FBAR of FIG. 7A.

FIG. 8 illustrates the structure of the BST-based FBAR device, accordingto still another embodiment of the present invention.

FIG. 9 illustrates the simulated behavior of a single BST-based FBARdevice in series and shunt configurations.

FIG. 10A illustrates a band pass filter circuit implemented using theBST-based FBAR devices according to one embodiment of the presentinvention.

FIG. 10B is a graph illustrating RF transmission measurements, as afunction of the frequency of the RF signal, of the band pass filtercircuit of FIG. 10A implemented using the BST-based FBAR devicesaccording to one embodiment of the present invention.

FIG. 10C is a graph illustrating how the RF transmission measurements ofthe band pass filter circuit of FIG. 10A implemented using the BST-basedFBAR devices change depending upon different DC bias voltages.

FIG. 11 illustrates a duplexer implemented using the BST-based FBARdevices according to one embodiment of the present invention.

FIG. 12A illustrates a conventional switched filter bank.

FIG. 12B illustrates a switched filter bank implemented using theBST-based FBAR devices according to one embodiment of the presentinvention.

DETAILED DESCRIPTION OF EMBODIMENTS

The Figures (FIG.) and the following description relate to preferredembodiments of the present invention by way of illustration only. Itshould be noted that from the following discussion, alternativeembodiments of the structures and methods disclosed herein will bereadily recognized as viable alternatives that may be employed withoutdeparting from the principles of the claimed invention.

Reference will now be made in detail to several embodiments of thepresent invention(s), examples of which are illustrated in theaccompanying figures. It is noted that wherever practicable similar orlike reference numbers may be used in the figures and may indicatesimilar or like functionality. The figures depict embodiments of thepresent invention for purposes of illustration only. One skilled in theart will readily recognize from the following description thatalternative embodiments of the structures and methods illustrated hereinmay be employed without departing from the principles of the inventiondescribed herein.

BST capacitors can be configured to exhibit the characteristics of anacoustic resonator if the BST capacitor is appropriately controlled. Asthe DC bias voltage to BST capacitors is increased, resonant dips in thereflection coefficient are observed. The frequency and depth of theresonance varies with the device area and electrode thicknesses as wellas the DC bias voltage. Such resonance in the BST varactors isconsistent with a thickness-mode acoustic resonance. BST (and alsoStrontium Titanate) exhibits a field-induced piezoelectricity, such thatunder bias the BST thin films can strongly couple electrical energy toacoustic vibration. This is sometimes called “electro-restrictive”behavior to distinguish the effect from a pure piezoelectric materialwhich would also exhibit the inverse effect (where a mechanicaldeformation leads to an electrical polarization). In the context of BSTvaractors, this resonance is not especially desirable since it iseffectively a loss mechanism that lowers the overall Q-factor of thedevice. For example, as the RF signal passes through a biased BSTcapacitor, part of its energy is converted into mechanical energy whichthen oscillates back-and-forth within the MIM structure forming astanding wave resonance. This standing wave will continue to draw energyaway from the RF signal as long as the DC electric field remains.Consequently, this effect results in transmission loss in the BSTcapacitors.

FIG. 2 is a graph illustrating RF transmission measurements, of thecapacitor 100 using the BST dielectric layer 120, as a function of thefrequency of the RF signal voltage. Two curves 210, 220 are shown,corresponding to different applied DC voltages. At zero applied DCvoltage, curve 210 shows a well-behaved flat response with nosignificant transmission loss. In contrast, at an applied DC voltage of20 V, curve 220 shows a large resonance and transmission loss appearingat a specific resonant frequency of about 3.7 GHz caused by thepiezoelectric effect of the BST material 120.

Such piezoelectric effects of the BST dielectric material 120, ifcarefully controlled, may be used to implement useful electricalcomponents. It is beneficial to control the piezoelectric effects of theBST dielectric material used in varactors to implement useful electroniccomponents.

FIG. 3 is a graph illustrating RF transmission measurements of the BSTcapacitor of FIG. 1 as a function of the frequency of the RF signalunder different DC bias voltages. The curve 310 shows the measurementsunder zero DC bias (V=0, off). At zero DC bias voltage, the BST-basedvaractor shows a well-behaved flat response with no significanttransmission loss, as illustrated in curve 310. In contrast, at anapplied DC voltage of 40 V, curve 315 shows a large anti-resonance andtransmission loss appearing at around 2.55 GHz. Also, at appliedvoltages of 10V, 20V, 30V, and 40V, the curves 320 show a largeresonance and transmission loss appearing at a certain frequency. Atsuch resonant frequency, the BST varactor is in the high-impedance,anti-resonance state leading to a deep notch in the frequency response.The specific frequencies at which the anti-resonance occur increases asthe DC bias voltage increases and the transmission loss at suchfrequency also increases as the DC bias voltages increase. In theoff-state, the insertion loss is a function of the capacitive reactanceof the BST varactor device. By increasing the device area, it ispossible to improve the insertion loss of the off-state and create amore attractive notch filter with a higher loaded Q-factor. Therefore,the BST varactor can be used as a switchable resonator that is switchedon with non-zero DC bias voltage but switched off with zero DC-biasvoltage. Further, the anti-resonant frequency and the transmissionlosses through the BST varactor can also be controlled using differentlevels of the DC bias voltage. In essence, the BST varactor functions asa switchable and adjustable FBAR (Film Bulk Acoustic Resonator) device.

FIG. 4 is a diagram of an equivalent circuit modeling a piezo-electrictransducer (acoustic resonator). Such equivalent circuit can also beused to model the piezo-electric characteristics of the BST varactor.The transducer can be represented by the KLM (Krimholtz, Leedhom, andMatthaei) model. The KLM model uses an equivalent transmission-linecircuit to model the one-dimensional acoustic wave problem. In the KLMmodel described in FIG. 4, V is the voltage applied to the acousticresonator, I is the current applied to the acoustic resonator, C_(c) isthe clamped capacitance, Z_(a) is the impedance looking into theacoustic resonator, 1:n is the turns ratio of a transformer thatconverts the electrical signal into acoustic resonance, v₁ and v₂ arethe particle velocities at the surfaces of the acoustic resonator, F₁and F₂ are the acoustic forces at the surfaces of the acousticresonator, Z₀ is the characteristic impedance of the acoustictransmission line, and γ is the propagation constant of the acoustictransmission line. In the KLM model, each acoustic layer (includingelectrodes, dielectrics, substrates, etc) is specified by a mass densityρ_(m) and an acoustic wave velocity ν_(p), from which equivalenttransmission-line parameters (the characteristic impedance Z₀ andpropagation constant β) can be computed as follows:

Z₀ = A ρ_(m)v_(p) $\beta = \frac{\omega}{v_{p}}$

where A is the active device area (electrode area) and ω is thefrequency of resonance. The acoustic loss in each layer is specified bya mechanical viscosity η, such that the attenuation factor α is given by

$\alpha = \frac{\eta \; \omega}{2\rho_{m}v_{p}}$

and each layer is defined by a complex propagation constant γ=α+jβ. Inaddition to these parameters, the piezoelectric layer (BST in thepresent invention) is further characterized by a piezoelectric strainconstant d_(m)[C/N] which relates the applied electric field to theresulting mechanical strain (deformation). It can be shown that theremaining equivalent circuit parameters in the KLM model are given by:

$n = {\frac{j\; \omega}{2h}\frac{Z_{0}}{\sinh \left( {{\gamma \; },2} \right)}}$$Z_{a} = {\frac{h^{2}}{\omega^{2}}\frac{\sinh \; \gamma \; l}{Z_{0}}}$

where

$h = \frac{c_{m}d_{m}}{ɛ^{S}}$ ɛ^(S) = ɛ − c_(m)d_(m)²

and l is the length of the acoustic transmission line, and c_(m) is thestiffness constant, related to the mass density and acoustic velocity asc_(m)=ρ_(m)ν_(p) ². Note that ε is the permittivity of the material withno mechanical stress, and ε^(S) is the permittivity that would bemeasured if the material were clamped to prevent deformation. Since thepiezoelectric strain constant often appears in combination with otherparameters, piezoelectric materials are often characterized by adimensionless piezoelectric coupling constant K², or theelectromechanical coupling constant k_(t) ², related by

$K^{2} = \frac{c_{m}d_{m}^{2}}{ɛ^{S}}$$k_{t}^{2} = \frac{K^{2}}{K^{2} + 1}$

These formulas are convenient for calculations related to complexmulti-layered structures.

FIG. 5 illustrates the structure and use of the BST-based FBAR 500according to one embodiment of the present invention. The BST-based FBARdevice 500 includes a BST (e.g., Ba_(x)Sr_(1-x)TiO₃) layer 520 disposedbetween a top electrode 510 a and a bottom electrode 510 b supported bya substrate 530 (e.g., Sapphire). The substrate material 530 is notlimited to Sapphire and other materials such as Silicon can be usedsuitably. The top electrode 510 a, the BST layer 520, the bottomelectrode 510 b, and the substrate 530 has a thickness of t₁, t₂, t₃,and t₄, respectively. As explained above, the BST-based FBAR device 500illustrates the characteristics of an FBAR device under non-zero DC biasvoltage. A voltage V_(g) 540 (including a DC component and an ACcomponent) is applied to the electrodes 510 a, 510 b through the inputimpedance Z_(g). A DC voltage generating an electric field of 1 MV/cmacross the FBAR device 500 can change the dielectric constant of the BSTmaterial 520 by factors of 2 to 3, leading to different frequencyresponses of the FBAR device 500 as illustrated in FIG. 4. However,under zero DC bias voltage, the FBAR device 500 loses thecharacteristics of an FBAR, and has characteristics similar to a simplethin film capacitor.

FIG. 6 illustrates the structure of the BST-based FBAR 600, according toone embodiment of the present invention. The FBAR 600 is fabricated on asapphire substrate 530, and includes top and bottom electrodes 510 a,510 b and a BST layer 520 disposed between the top and bottom electrodes510 a, 510 b. The substrate material 530 is not limited to Sapphire andother materials such as Silicon can be used suitably. Note that the FBAR600 device is formed over air 620 such that there is an air gap 620between the bottom electrode 510 b and the sapphire substrate 530. Theair gap 620 reduces the damping of the resonance caused by the substrate530. DC bias 650 is applied to the top electrode 510 a, and the bottomelectrode 510 b is connected to DC ground 651. The RF signal 652 isinput to the top electrode 510 a, passes through the BST layer 520, andis output 654 from the bottom electrode 510 b.

FIG. 7A illustrates the structure of the BST-based FBAR 700, accordingto another embodiment of the present invention. The FBAR 700 isfabricated on a sapphire substrate 530, and includes top and bottomelectrodes 510 a, 510 b and a BST layer 520 disposed between the top andbottom electrodes 510 a, 510 b. The substrate material 530 is notlimited to Sapphire and other materials such as Silicon can be usedsuitably. Note that the FBAR 700 device in FIG. 7A is fabricated on anacoustic reflector 705 which is disposed on the sapphire substrate 530.The acoustic reflector 705 (also referred to as an “acoustic mirror”)functions to reduce the damping of the resonance caused by the substrate530, as will be explained in more detail with reference to FIG. 7B. DCbias 780 is applied to the top electrode 510 a, and the bottom electrode510 b is connected to DC ground 781. The RF signal 782 is input to thetop electrode 510 a, passes through the BST layer 520, and is output 784from the bottom electrode 510 b.

FIG. 7B illustrates the various structures of the acoustic reflector 705that can be used in the BST-based FBAR 700 of FIG. 7A. The acousticreflector 705 is made from alternating quarter-wavelength layers of highand low acoustic impedance materials. The acoustic impedance is relatedto the mass-density and sound velocity of the materials. For a BST-baseddevice, platinum (Pt) layers are highly desirable because Pt isrefractory (can withstand high BST deposition temperatures), resistantto oxidation, and has a large work function, forming a large Schottkybarrier at the interface to reduce leakage. Pt also has an extremelylarge mass density, making it very attractive as the high impedancelayer in an acoustic mirror stack. At the opposite end of the spectrum,SiO₂ is a very attractive and commonly-used as a low-impedance layer.FIG. 7B shows the improvement in reflection coefficient for three simpleacoustic mirror stacks 750, 760, 770. The acoustic reflector 750 has asimple Pt electrode disposed on a sapphire substrate, showing areflection coefficient as in graph 752. The acoustic reflector 760 has a2-layer Pt/SiO₂ mirror disposed on a sapphire substrate, showing areflection coefficient as in graph 762. The acoustic reflector 770 has a4-layer Pt/SiO₂ mirror disposed on a sapphire substrate, showing areflection coefficient as in graph 772. In the latter two cases(acoustic mirrors 760, 770) the layer thicknesses were chosen to be aquarter-wavelength at 2.5 GHz. It can be seen the reflection coefficientimproves as the acoustic reflector has multiple Pt/SiO₂ layers as in theacoustic mirror 760, 770.

FIG. 8 illustrates the structure of the BST-based FBAR device 800,according to still another embodiment of the present invention. The FBAR800 is fabricated on a sapphire substrate 830, and includes top andbottom electrodes 510 a, 510 b and a BST layer 520 disposed between thetop and bottom electrodes 510 a, 510 b. The substrate material 530 isnot limited to Sapphire and other materials such as Silicon can be usedsuitably. Note that the sapphire substrate 830 has a pair of taperedparts (tapered as they become further away from the bottom electrode 510b) with an air gap 820 disposed between the pair of tapered parts of thesapphire substrate 830. Thus, the FBAR 800 device in FIG. 8 is disposedpartly on the sapphire substrate 830 and partly on the air gap 820. Theair gap 820 is created by milling away the substrate 530. The air gap820 also reduces the damping of the resonance of the FBAR device 800that would otherwise be caused by the substrate 530. The via 860provides a conduit for electrical connection to the bottom electrode 510b. DC bias 850 is applied to the top electrode 510 a, and the bottomelectrode 510 b is connected to DC ground 851. The RF signal 852 isinput to the top electrode 510 a, passes through the BST layer 520, andis output 854 from the bottom electrode 510 b.

FIG. 9 illustrates the simulated behavior of a single BST-based FBARdevice in series 900 and shunt 950 configurations. A clear way toexploit the voltage-dependent piezoelectric coupling of the BST-basedFBAR devices is by designing the circuits to maximize the impedancedifferential between “off” (zero bias) and “on” (maximum bias) states.The series configuration 900 includes the FBAR device 902 in series withan input impedance 904 (Z₀) receiving an input RF input signal 908 andan output impedance 906 (Z₀) for the RF output signal. In thetransmission loss graph 920 for the series configuration 900, it can beseen that the anti-resonance occurs around 2.55 GHz when a DC bias of 40V is applied to the FBAR device 902, while no anti-resonance occurs whenno DC bias is applied (0 Volt). The shunt configuration 950 includes aninput impedance 918 (Z₀) receiving an RF input signal 918, and the FBARdevice 912 in parallel with the output impedance 916 (Z₀) for the RFoutput signal. In the transmission loss graph 940 for the shuntconfiguration 950, it can be seen that the anti-resonance also occursaround 2.55 GHz when a DC bias of 40 V is applied to the FBAR device912, while no anti-resonance occurs when no DC bias is applied (0 Volt).Note that the simulations in FIG. 9 include electrical losses associatedwith the material loss tangent and series resistance of the electrodes,as well as mechanical damping and acoustic losses in the electrodes andsubstrates of the FBAR devices 902, 912. Further improvements inperformance can be obtained by using more advanced acoustic mirrorstacks and higher-quality BST films. BST films are usually engineeredfor high-tunability at the expense of higher loss tangents; in thisapplication, BST composition and deposition conditions could beoptimized purely for low loss tangents, since a large capacitivetunability is not necessary for the proper functioning of these devices.

The switched and tunable resonance properties of the BST-based FBARdevices can be used in modern communication systems wherefrequency-agile or reconfigurable components are becoming increasinglyimportant and necessary to cope with a multitude of signal frequenciesand modulation formats, including analog front-end components near theantenna, such as filters, duplexers, antenna and amplifier matchingnetworks, etc. This is a difficult problem area for electronics becausethe transmit power levels and associated RF voltage swings can be quitelarge, raising breakdown and linearity concerns, while at the same timethe receive signal levels are extremely low, placing a premium oninsertion loss to maintain an acceptable signal-to-noise ratio. Sincethe BST-based FBAR devices according to the present invention areswitchable and tunable, they are functionally equivalent to ahigh-selectivity filter and a low-loss switch, where the switch draws noDC power. An array of such filters could implement a very compact andreconfigurable high-selectivity filter bank. Similarly, BST-based FBARscould be combined to make a low-loss and high-selectivity duplexer or TRswitch. There are many other possibilities for frequency-agilecomponents.

FIG. 10A illustrates a band pass filter circuit implemented using theBST-based FBAR devices according to one embodiment of the presentinvention. The “ladder” filter network shown in FIG. 10A includescapacitors 1004, 1006, 1008 and BST-based FBAR devices 1010, 1012,receiving an RF input signal through an input impedance 1002 andgenerating an RF output signal across the output impedance 1014. TheBST-based FBAR devices 1010, 1012 may be of any configuration asillustrated in FIG. 5, 6, 7A, or 8. The “ladder” filter network shown inFIG. 10A is essentially a combination of capacitively-coupled shuntresonators, and can be optimized using classical filter techniques.

FIG. 10B is a graph illustrating RF transmission simulation results, asa function of the frequency of the RF signal, of the band pass filtercircuit implemented using the BST-based FBAR devices according to oneembodiment of the present invention in FIG. 10A. It is apparent that thefilter network of FIG. 10A exhibits the characteristics of a 2-pole bandpass filter when a DC bias of 40 V is applied to the FBAR devices 1010,1012 but that a steady frequency response occurs when the DC bias is off(0 volt). In this simple filter network with only two BST FBAR devices1010, 1012, over a 40 dB on-off dynamic range can be obtained, with lessthan 3 dB insertion loss.

FIG. 10C is a graph illustrating how the RF transmission measurements ofthe band pass filter circuit implemented using the BST-based FBARdevices as in FIG. 10A change depending upon different DC bias voltagesapplied to the BST-based FBAR devices. When the DC bias voltage appliedto the BST-based FBAR devices 1010, 1012 is V1, the center frequency ofthe band pass filter is positioned at F1. By changing the DC voltageapplied to the BST-based FBAR devices 1010, 1012 to V2, the filterresponse shifts upwards and settles at the new center frequency of F2.Therefore, the filter of FIG. 10A is not only switchable based onwhether a DC bias voltage is applied, but its frequency response is alsotunable based on the level of the DC bias voltage. Thus, the filter ofFIG. 10A essentially has multiple frequency bands. For example, a DCbias voltage of V1 applied to the filter network would enable operationof Band 1 centered at F1 while suppressing Band 2 centered at V2.Conversely, a DC bias voltage of V2 would enable Band 2 centered at F2while disabling Band 1 centered at F1.

With building blocks such as the filter circuit in FIG. 10A, numeroushigh-level functional components can be realized. FIG. 11 illustrates aduplexer 1100 implemented using the BST-based FBAR devices according toone embodiment of the present invention. The duplexer 1100 includes twoBST-based FBAR filter circuits 1106, 1008, such as those shown in FIG.10A, to route signals between the antenna 1110 and the transmit (Tx)block 1102 or the receive (Rx) block 1104. When the BST-based FBARfilter 1106 in the transmit path is activated by applying a non-zero DCbias voltage (V_(switch)) to the BST-based FBAR 1106, the BST-based FBARfilter 1108 in the receive path is turned off, thus isolating thereceiver 1104 from the transmit signal 1102. When the BST-based FBARfilter 1108 in the receive path is activated by applying a non-zero DCbias voltage (˜V_(switch)) to the BST-based FBAR 1108, the BST-basedFBAR filter 1106 in the transmit path is turned off, thus isolating thetransmitter 1102 from the receive signal 1104. The great advantage hereis that both the switching and filtering functions can be integratedtogether in one device, which is monolithically integrated on a singlechip, and furthermore the DC control circuit draws no current. Aduplexer functionality can be implemented using filters with differentpass band frequencies for the Tx and Rx paths, by applying different DCbias voltages to the BST-based FBAR filters 1106, 1108.

Voltage-selectable band-pass or band-reject filter structures like thoseshown in FIG. 10A can be combined in parallel to implement compact,reconfigurable filter banks FIG. 12A illustrates a conventional switchedfilter bank 1200, including conventional FBAR switches 1202, 1204. Theconventional FBAR switches 1202, 1204 are not independently switchable,and thus the DC switching signals (V_(switch) and ˜V_(switch)) areapplied to the switches 1210, 1212, 1214, 1216 to externally turn on oroff the conventional FBAR filters 1202, 1204.

In contrast, FIG. 12B illustrates a switched filter bank 1250implemented using the BST-based FBAR devices according to one embodimentof the present invention. The filter bank 1250 includes two BST-basedFBAR filters 1252, 1254. Note that the BST-based FBAR filters 1252, 1254are switched directly by the DC switching signals (V_(switch) and˜V_(switch)) because the BST-based FBAR filters 1252, 1254 areswitchable based on whether a DC bias signal is applied thereto. Thus,the switched filter bank 1250 does not require external switches toswitch on or off the BST-based FBAR filters 1252, 1254.

Upon reading this disclosure, those of ordinary skill in the art willappreciate still additional alternative structural and functionaldesigns for a BST-based FBAR device and its applications through thedisclosed principles of the present invention. Thus, while particularembodiments and applications of the present invention have beenillustrated and described, it is to be understood that the invention isnot limited to the precise construction and components disclosed herein.Various modifications, changes and variations which will be apparent tothose skilled in the art may be made in the arrangement, operation anddetails of the method and apparatus of the present invention disclosedherein without departing from the spirit and scope of the invention asdefined in the appended claims.

1. An acoustic resonator comprising: a first electrode; a secondelectrode; a barium strontium titanate (BST) dielectric layer disposedbetween the first electrode and the second electrode, the acousticresonator being configured to be switched on with a resonant frequencyby applying a DC (direct current) bias voltage across the BST dielectriclayer, and the acoustic resonator being configured to be switched off byremoving the DC bias voltage across the BST dielectric layer.
 2. Theacoustic resonator of claim 1, wherein the resonant frequency is tunedbased on a level of the DC bias voltage.
 3. The acoustic resonator ofclaim 2, wherein the resonant frequency increases as the level of the DCbias voltage increases.
 4. The acoustic resonator of claim 1, whereinthe acoustic resonator is formed on a sapphire substrate.
 5. Theacoustic resonator of claim 1, wherein the acoustic resonator is formedover an air gap disposed between the second electrode and a substratesupporting the acoustic resonator.
 6. The acoustic resonator of claim 1,wherein the acoustic resonator is formed over an acoustic reflectordisposed between the second electrode and a substrate supporting theacoustic resonator, the acoustic reflector reducing damping of resonanceof the acoustic resonator caused by the substrate.
 7. The acousticresonator of claim 6, wherein the acoustic reflector comprises aPlatinum (Pt) layer.
 8. The acoustic resonator of claim 6, wherein theacoustic reflector comprises a plurality of alternating layers ofPlatinum (Pt) and Silicon Dioxide (SiO₂).
 9. The acoustic resonator ofclaim 1, wherein the acoustic resonator includes a first part formedover a substrate and a second part formed over an air gap.
 10. Theacoustic resonator of claim 1, wherein the acoustic resonator is used aspart of a switchable filter.
 11. The acoustic resonator of claim 1,wherein the acoustic resonator is used as part of a duplexer fortransmitting and receiving a radio frequency signal over an antenna. 12.A filter for filtering a frequency range of a signal, the filtercomprising: at least a capacitor; and at least an acoustic resonatorcoupled to the capacitor, the acoustic resonator including: a firstelectrode; a second electrode; a barium strontium titanate (BST)dielectric layer disposed between the first electrode and the secondelectrode, the acoustic resonator being configured to be switched onwith a resonant frequency by applying a DC (direct current) bias voltageacross the BST dielectric layer, and the acoustic resonator beingconfigured to be switched off by removing the DC bias voltage across theBST dielectric layer.
 13. The filter of claim 12, wherein the resonantfrequency is tuned based on a level of the DC bias voltage.
 14. Thefilter of claim 13, wherein the resonant frequency increases as thelevel of the DC bias voltage increases.
 15. The filter of claim 12,wherein the acoustic resonator is formed on a sapphire substrate. 16.The filter of claim 12, wherein the acoustic resonator is formed over anair gap disposed between the second electrode and a substrate supportingthe acoustic resonator.
 17. The filter of claim 12, wherein the acousticresonator is formed over an acoustic reflector disposed between thesecond electrode and a substrate supporting the acoustic resonator, theacoustic reflector reducing damping of resonance of the acousticresonator caused by the substrate.
 18. The filter of claim 17, whereinthe acoustic reflector comprises a Platinum (Pt) layer.
 19. The filterof claim 17, wherein the acoustic reflector comprises a plurality ofalternating layers of Platinum (Pt) and Silicon Dioxide (SiO₂).
 20. Thefilter of claim 12, wherein the acoustic resonator includes a first partformed over a substrate and a second part formed over an air gap.